1. Field of the Invention
This invention relates to an improved single-stage, single-switch, input-current-shaping technique with fast output-voltage regulation and reduced switching losses and, more particularly, to the single-stage, flyback input current-shaping circuit, which operates at the boundary of the discontinuous and continuous magnetizing current of the transformer in the entire line and load ranges.
2. Description of the Prior Art
The harmonic content of the line current drawn from the ac mains by a piece of electronic equipment is regulated by a number of standards. To comply with these standards, input-current correction (PFC) (also referred to in the art as power factor correction) of off-line power supplies is necessary. So far, a variety of passive and active ICS techniques have been proposed. While the passive techniques can be the best choice in many cost-sensitive applications, the active ICS techniques are used in the majority of applications due to their superior performance.
The most commonly used active approach that meets high power-quality requirements is the "two-stage" approach. In this approach, a non-isolated boost-like converter, which is controlled so that the rectified line current follows the rectified line voltage, is used as the input stage that creates an intermediate dc bus with a relatively large second-harmonic ripple. This ICS stage is then followed by a dc/dc converter which provides isolation and high-bandwidth voltage regulation. For high-power levels, the ICS stage is operated in the continuous-conduction mode (CCM), while the discontinuous-conduction-mode (DCM) operation is commonly used at lower power levels due to a simpler control.
In an effort to reduce the component count and also improve the performance, a number of "single-stage" ICS techniques have been introduced recently. In a single-stage approach, input-current shaping, isolation, and high-bandwidth control are performed in a single step, i.e., without creating an intermediate dc bus. Generally, these converters use an internal energy-storage capacitor to handle the differences between the varying instantaneous input power and a constant output power.
Among the single-stage circuits, a number of circuits described in M. M. Jovanovic and L. Huber, "Single-Stage, Single-Switch Isolated Power-Supply Technique with Input-Current Shaping and Fast Output-Voltage Regulation," patent application Ser. No. 08/669,001 filed on Jun. 21, 1996; F. S. Tsai, et al., "Low Cost AC-to-DC Converter Having Input Current with Reduced Harmonics," U.S. Pat. No. 5,652,700; and J. Qian and F. C. Lee, "A High Efficient Single Stage Single Switch High Power Factor AC/DC Converter with Universal Input," IEEE Applied Power Electronics Conference (APEC) Proc., pp. 281-287, February 1997, seem particularly attractive because they can be implemented with only one semiconductor switch and a simple control. All these single-stage, single-switch input-current shapers (S.sup.4 ICS) integrate the boost-converter front end with the forward-converter or the flyback-converter dc/dc stage. FIG. 1 shows the S.sup.4 ICS flyback converter implementation introduced in the Ser. No. 08/669,001 application. Transformer windings N.sub.1 and N.sub.2 are used to reduce the voltage of the energy-storage capacitor C.sub.B. In addition, winding N.sub.2 is utilized for the direct energy transfer from the input to the output. A small input capacitor C.sub.in is used to filter out the switching-frequency ripple of the ICS-inductor current. Consequently, the rectified line current is the average of the current flowing from ICS inductor L.sub.B.
One side of boost inductor L.sub.B is coupled to full-wave rectified input v.sub.in(rec) derived from the ac mains. Winding N.sub.2 of isolation transformer T.sub.1 is coupled, via diode D.sub.2, between the other side of boost inductor L.sub.B and one side of primary winding N.sub.P of isolation transformer T.sub.1. Winding N.sub.1 of isolation transformer T.sub.1 is coupled, via diode D.sub.1, between the other side of boost inductor L.sub.B and the other side of primary winding N.sub.P.
As explained in the Ser. No. 08/669,001 application, in the circuit in FIG. 1, two additional primary windings N.sub.1 and N.sub.2 are employed to keep the voltage of energy-storage (bulk) capacitor C.sub.B below the desired level of 450 V in the entire line and load ranges. Winding N.sub.1 appears in series with boost inductor L.sub.B during the on-time of switch SW, whereas winding N.sub.2 appears in series with L.sub.B during the off-time of the switch. By connecting the windings so that the voltages across them when they conduct the boost-inductor current are in the opposition to the line voltage, the volt-second balance of the boost-inductor core is achieved at a substantially lower voltage of the energy-storage capacitor compared to the corresponding circuit without the windings. In addition, in the circuit in FIG. 1, winding N.sub.2 provides a path for a direct transfer of a part of the input energy to the output during the off-time of the switch. For a properly selected number of turns of winding N.sub.2, this direct energy transfer increases the conversion efficiency of the circuit.
As explained in the Ser. No. 08/669,001 application, the S.sup.4 ICS flyback converter in FIG. 1 can operate either with a discontinuous or a continuous current of boost inductor L.sub.B. Generally, the continuous-conduction mode (CCM) of operation offers a slightly higher efficiency compared to the discontinuous-conduction mode (DCM). However, the DCM operation gives a lower total harmonic distortion (THD) of the line current compared to that of the CCM operation.
The output-voltage regulation of the S.sup.4 ICS flyback converter in FIG. 1, is implemented by a constant-frequency two-loop control. The two loops, current loop T.sub.i and voltage loop T.sub.v, are indicated in FIG. 1. In the voltage loop, output voltage V.sub.o is scaled down with resistive divider R.sub.1 -R.sub.2 before it is compared to reference voltage V.sub.REF at the input of error amplifier EA. The output of error amplifier EA, whose voltage V.sub.EA is proportional to the error between output voltage V.sub.o and reference voltage V.sub.REF, is connected to the inverting input of the comparator. In the current loop, the current of switch SW is sensed during the switch on-time, and converted to a suitable voltage signal by sensing device (resistor) R.sub.i before it is connected to the non-inverting input of the comparator. In the control circuit in FIG. 1, a constant-frequency clock signal initiates the turn-on of switch SW by setting the RS latch output Q high. Since during the time period that switch SW is closed, switch current i.sub.SW increases, the sensed voltage R.sub.i i.sub.SW at the non-inverting input of the comparator also increases. When voltage R.sub.i i.sub.SW reaches the V.sub.EA voltage level, the output of the comparator transitions from the low to the high state resetting the RS latch and turning off switch SW. In this regulation scheme, output voltage of the error amplifier V.sub.EA automatically adjusts to the level which is necessary to produce a duty cycle of switch SW that is required to maintain output voltage V.sub.o constant.
Generally, in ICS application, the rectified line voltage, which is the input voltage to the converter, contains a large ripple. This input-voltage ripple propagates through the power stage causing an increased output-voltage ripple at the rectified-line frequency. To eliminate the rectified-line-voltage component of the output-voltage ripple, it is necessary to design the output-voltage feedback loop (T.sub.v) with a bandwidth which is wide enough to attenuate the ripple to the desired value. The desired bandwidth, regulation accuracy, and control-loop stability are set by a proper selection of compensation impedances Z.sub.1 and Z.sub.2, shown in FIG. 1. It should be noted that current loop T.sub.i in FIG. 1 also plays a major roll in the attenuation of the output-voltage ripple.
FIG. 2 shows the key waveforms of the S.sup.4 ICS flyback converter in FIG. 1 operating with a discontinuous boost inductor current i.sub.LB and with a continuous magnetizing current i.sub.M of the flyback transformer. To facilitate the explanation of operation, the transformer in FIG. 1 is shown as the parallel combination of transformer's magnetizing inductance L.sub.M and the ideal transformer consisting of primary winding N.sub.P and secondary winding N.sub.S. Since L.sub.B works in the DCM, i.sub.LB is zero prior to the turn-on of switch SW at t=T.sub.0. As can be seen from FIG. 2, after clock initiates the turn-on of switch SW at t=T.sub.0, boost inductor current i.sub.LB starts increasing linearly with a slope of (V.sub.in(rec) -(N.sub.1 /N.sub.P)V.sub.B)/L.sub.B. At the same time, due to a positive voltage across the primary winding of the transformer, secondary current i.sub.S cannot flow because rectifier D.sub.F is reverse biased. Since the primary voltage during the on-time is constant and equal to the storage-capacitor voltage V.sub.B, magnetizing current i.sub.M increases with a constant slope of V.sub.B /L.sub.M, where L.sub.M is the primary-side-referred magnetizing inductance of the transformer. Also, because of the magnetic coupling between windings N.sub.1 and N.sub.P, current i.sub.LB, which flows through winding N.sub.1 during the on-time of switch SW, induces current i.sub.P1 =-(N.sub.1 /N.sub.P)i.sub.LB in the primary winding of the transformer. Due to the existence of negative current i.sub.P1, the component of magnetizing current i.sub.M supplied from energy-storage capacitor C.sub.B, i.sub.CB, is reduced. Since during the on-time switch current is i.sub.SW is the sum of boost-inductor current i.sub.LB and primary current i.sub.P =i.sub.M +i.sub.P1 =i.sub.CB, i.sub.SW also increases linearly, as shown in FIG. 2. The conduction of switch SW is terminated at t=T.sub.1 when sensed voltage R.sub.i i.sub.SW at the non-inverting input of the comparator reaches the level of error-amplifier output voltage V.sub.EA at the inverting input of the comparator. After switch SW is turned off, boost inductor current i.sub.LB is diverted from winding N.sub.1 to winding N.sub.2 forcing the conduction of diode D.sub.2. At the same time, due to a positive voltage on the secondary winding, diode D.sub.F starts conducting secondary current i.sub.S, while primary current i.sub.P ceases flowing. It should be also noted that because of the magnetic coupling between windings N.sub.2 and N.sub.S, during the off-time a part of the input energy is directly transferred to the output instead of first being stored in energy-storage capacitor C.sub.B. This direct energy transfer occurs as long as decreasing boost inductor current i.sub.LB is flowing through winding N.sub.2. When i.sub.LB becomes zero at t=T.sub.2, diode D.sub.2 stops conducting, and the entire secondary current i.sub.S consists of the decreasing magnetizing current.
From v.sub.SW waveform in FIG. 2, it can be seen that during the off time, the voltage across switch SW is given by V.sub.SW(off) =V.sub.B +nV.sub.o, where n=N.sub.P /N.sub.S is the turns ratio of the transformer. As a result, the energy stored in the parasitic output capacitance of the switch, C.sub.oss, prior to its turn on is equal to ##EQU1##
This energy is dissipated in the switch when the switch is turned on. If the circuit in FIG. 1 operates at constant switching frequency f.sub.S, the power dissipation of the switch associated with the capacitive-discharge turn-on loss is ##EQU2##
As can be seen from Eq. (2), capacitive-discharge turn-on switching loss P.sub.ON(cap) increases linearly with the switching frequency, f.sub.S, and quadratically with the voltage across the switch immediately before the switch turns on, V.sub.SW(off). Since V.sub.SW(off) =V.sub.B +nV.sub.o increases as the line voltage increases because V.sub.B increases with the line voltage, the capacitive-discharge turn-on switching loss is maximum at high line. Generally, MOSFET switches with lower on-resistances R.sub.DS(on), which reduce conduction losses, possess larger output capacitances C.sub.oss. Therefore, at high switching frequencies, the capacitive-discharge turn-on switching loss has a detrimental effect on the efficiency of the circuit in FIG. 1, especially, at high line. Because P.sub.ON(cap) does not depend on the load current but only on the line voltage, P.sub.ON(cap) dominates the switch loss at light loads and, consequently, limits the light-load efficiency. A reduced light-load efficiency makes very difficult to comply with Environmental Protection Agency's Energy Star requirement which sets a voluntary power-consumption limit of an idling personal computer to 60 W (30 W for the monitor and 30 W for the computer box).
Similar conclusions with respect to the capacitive-discharge turn-on loss can be drawn for the S.sup.4 ICS flyback converter in FIG. 1 operating with a discontinuous boost inductor current i.sub.LB and with a discontinuous magnetizing current i.sub.M of the flyback transformer. As shown in FIG. 3, the voltage of switch SW during the off-time of the switch in the S.sup.4 ICS flyback converter operating with a discontinuous magnetizing current of the transformer is equal to V.sub.SW(off) =V.sub.B, if the switch capacitance is negligible. Otherwise, V.sub.SW(off) is oscillating around V.sub.B with the amplitude equal to nV.sub.o because of the resonance between the magnetizing inductance of the transformer, L.sub.M, and the output capacitance of the switch, C.sub.oss. Therefore, the S.sup.4 ICS flyback converter operating with a discontinuous magnetizing current of the transformer also suffers from capacitive-discharge turn-on switching loss P.sub.ON(cap). Generally, this loss is dependent on the turn-on switching instant due to the resonant nature of the switch voltage during the off-time. However, the worst-case P.sub.ON(cap) in the S.sup.4 ICS flyback converter operating with a discontinuous magnetizing current of the transformer (waveforms shown in FIG. 3) is the same to that of the converter operating with a continuous magnetizing current of the transformer (waveforms shown in FIG. 2). Finally, it should be noted that the mode of operation of boost inductor L.sub.B (DCM or CCM) does not have any effect on the P.sub.ON(cap).